This invention relates to a high temperature superconductor (HTS) tunable filter. More particularly, this invention relates to an HTS filter tunable by variation of a microelectromechanical capacitor.
The need for a high-quality factor (Q), low insertion loss tunable filter pervades a wide range of microwave and RF applications, in both the military, e.g., RADAR, communications and electronic intelligence (ELINT), and the commercial fields such as in various communications applications, including cellular. Placing a sharply defined bandpass filter directly at the receiver antenna input will often eliminate various adverse effects resulting from strong interfering signals at frequencies near the desired signal frequency in such applications. Because of the location of the filter at the receiver antenna input, the insertion loss must be very low to not degrade the noise figure. In most filter technologies, achieving a low insertion loss requires a corresponding compromise in filter steepness or selectivity. In the present invention, the extremely low loss property of high-temperature superconductor (HTS) filter elements provides an attractive solution, achieving a very low insertion loss yet simultaneously allowing a high selectivity/steepness bandpass definition.
In many applications, particularly where frequency hopping is used, a receiver filter must be tunable to either to select a desired frequency or to trap an interfering signal frequency. Thus, the insertion of a linear tunable filter between the receiver antenna and the first nonlinear element (typically a low-noise amplifier or mixer) in the receiver offers, providing that the insertion loss is very low, substantial advantages in a wide range of RF and microwave systems. For example, in RADAR systems, high amplitude interfering signals, either from xe2x80x9cfriendlyxe2x80x9d nearby sources, or from jammers, can desensitize receivers or intermodulate with high-amplitude clutter signal levels to give false target indications. In high-density signal environments, RADAR warning systems frequently become completely unusable.
Both lumped element and distributed element filters suffer from these and other problems. For example, while distributed-element YIG-tuned filters have been used, the high level of insertion loss (usually greater than 10 dB) of suitable YIG filters necessitates their use on a xe2x80x9cswitch in when absolutely necessaryxe2x80x9d basis only, as the degradation to noise figure would generally be unacceptable. Lumped element filters also suffer problems. For a lumped element filter to be tunable, the filter requires either a tunable capacitor, or a tunable inductive element. The vast majority of RF tunable lumped element filters have used varactor diodes. Such a design amounts to using a tunable capacitor because varactor diodes, through changing the reverse bias voltage, vary the depletion thickness and hence the PN junction capacitance. While varactors are simple and robust, they have limited Q""s, and suffer from the problem that the linear process that tunes them extends all of the way to the signal frequency, so that high-amplitude signals create, through the resulting nonlinearities, undesirable intermodulate products, etc. The same problems of poor Q and high-frequency nonlinearities are anticipated for xe2x80x9ctunable materialsxe2x80x9d such as ferroelectrics.
Consider the case of a conventional varactor diode. In a varactor, the motion of electrons accomplishes the tuning itself. As the reverse bias (Vr) on the junction of the varactor is changed, then in accordance with Poisson""s Equation, the width of the PN junction depletion region changes which alters the capacitance of the junction (Cj). Because the tuning mechanism of varactors is electronic, the tuning speed is extremely fast. Unfortunately, this also leads to a serious associated disadvantage: limited dynamic range. Because the Cj (Vr) relationship is nearly instantaneous in response, extending to changes in Vr at the signal frequency itself, and because the signal (frequently in a resonantly magnified form) appears as a component of the junction bias voltage, Vr, the signal itself parametrically modulates the junction capacitance. If the signal amplitude across the varactor is very small in comparison to the dc bias, the effect is not too serious. Unfortunately, for high signal amplitudes, this parametric modulation of the capacitance can produce severe cross-modulation (IM) effects between different signals, as well as harmonic generation and other undesirable effects. While these signal-frequency varactor capacitance variations are the basis of useful devices such as parametric amplifiers, subharmonic oscillators, frequency multipliers, and many other useful microwave circuits, in the signal paths of conventional receivers they are an anathema. This inherent intermodulation or dynamic range problem will presumably extend to xe2x80x9ctunable materialsxe2x80x9d, such as ferroelectrics or other materials in which the change of dielectric constant (∈r) with applied electric field (E) is exploited to tune a circuit. As long as the ∈r (E) relationship applies out to the signal frequency, then the presence of the signal as a component of the E will lead to the same intermodulation problems that the varactors have.
In addition to the intermodulation/dynamic range problems of varactors, these conventional tuning devices also have serious limitations in Q, or tuning selectivity. Because the varactors operate by varying the depletion region width of a P-N junction, this means that at lower reverse biases (higher capacitances), there is a substantial amount of undepleted moderately-doped semiconductor material between the contacts and the junction that offers significant series resistance (Rac) to ac current flow. Since the Q of a varactor of junction capacitance Cj and series resistance Rac at the signal frequency f is given by Q=1/(2 f Cj Rac), this means that the varactor Q values are limited, particularly at higher frequencies. For example, a typical commercial varactor might have Cj=2.35 pF with Rac=1.0 xcexa9 at Vr=xe2x88x924V, or Cj=1.70 pF with Rac=0.82 xcexa9 at Vr=xe2x88x9210V,corresponding to Q values at f=1.0 GHz of Q=68 at Vr=xe2x88x924V or Q=114 at Vr=xe2x88x9210V (or f=10.0 GHz values of Q=6.8 and Q=11.4, respectively). Considering that an interesting X-band (f=10 GHz) RADAR application might want a bandwidth of f=20 MHz (the full width at half-maximum or FWHM), corresponding to a Q=500 quality factor, we see that available varactors have inadequate Q (too much loss) to meet such requirements. While the mechanisms are different, this will very likely apply to the use of ferroelectrics or other xe2x80x9ctunable materialsxe2x80x9d. A general characteristic of materials which exhibit the field-dependent dielectric constant nonlinearities (that makes them tunable) is that they exhibit substantial values of the imaginary part of the dielectric constant (or equivalently, loss tangent). This makes it unlikely that, as in varactors, these xe2x80x9ctunable materialsxe2x80x9d will be capable of achieving high Q""s, particularly at high signal frequencies.
An additional problem with both varactors and xe2x80x9ctunable materialsxe2x80x9d for circuits with high values of Q is that these are basically two-terminal devices; that is, the dc tuning voltage must be applied between the same two electrodes to which the signal voltage is applied. The standard technique is to apply the dc tuning bias through a xe2x80x9cbias teexe2x80x9d-like circuit designed to represent a high reactive impedance to the signal frequency to prevent loss of signal power out the bias port (as this would effectively reduce the Q). However, while the design of bias circuits that limit the loss of energy to a percent, or a fraction of a percent, even losses of a fraction of a percent are not nearly good enough for very high Q circuits (e.g., Q""s in the 103 to  greater than 104 range, as achievable with HTS resonators). It would be much easier to design such very high Q circuits using three-terminal, or preferably 4-terminal (two-port) variable capacitors in which the tuning voltage is applied to a completely different pair of electrodes from those across which the signal voltage is applied (with an inherent high degree of isolation between the signal and bias ports).
One new form of variable capacitor which avoids the problems of varactors or xe2x80x9ctunable materialsxe2x80x9d approaches is the microelectromechanical (MEMS) variable capacitor such as that disclosed by U.S. Pat. No. 5,696,662. A number of MEMS variable capacitor device structures have been proposed, including elaborate lateral-motion interdigitated electrode capacitor structures. In the simple vertical motion, parallel plate form of this device, a thin layer of dielectric separating normal metal plates (or a normal metal plate from very heavily doped silicon) is etched out in processing to leave a very narrow gap between the plates. The thin top plate is suspended on four highly compliant thin beams which terminate on posts (regions under which the spacer dielectric has not been removed). The device is ordinarily operated in an evacuated package to allow substantial voltages to be applied across the narrow gap between plates without air breakdown (and to eliminate air effects on motion of the plate and noise). When a dc tuning voltage is applied between the plates, the small electrostatic attractive force, due to the high compliance of the support beams, causes substantial deflection of the movable plate toward the fixed plate or substrate, increasing the capacitance.
Because the change of capacitance, at least in the metal-to-metal plate version of the MEMS variable capacitor, is due entirely to mechanical motion of the plate (as opposed to xe2x80x9cinstantaneousxe2x80x9d electronic motion effects as in varactors or xe2x80x9ctunable materialsxe2x80x9d), the frequency response is limited by the plate mass to far below signal frequencies of interest. Consequently, these MEMS devices will be free of measurable intermodulation or harmonic distortion effects, or other dynamic range problems (up to the point where the combination of bias plus signal voltage across the narrow gap between plates begins to lead to nonlinear current leakage or breakdown effects).
In addition to their freedom from IM/dynamic range problems, normal metal plate MEMS variable capacitor structures offer the potential for substantially lower losses and higher Q""s. While the simple parallel plate MEMS structure, which uses the top and bottom plates as the two capacitor electrodes, has a Q problem due to the skin effect resistance, Rac, of the long narrow metal leads down the compliant beams supporting the movable plate, an alternative structure is possible which avoids this problem. If the top (movable) plate is made electrically xe2x80x9cfloatingxe2x80x9d (from a signal standpoint; it would still have a dc bias lead on it), and the fixed bottom plate split into two equal parts, these two split plates can be used as the signal leads to the MEMS variable capacitor. (The capacitance value is halved, of course, but the tuning range is preserved.) In this xe2x80x9cfloating platexe2x80x9d configuration, passage of ac current through the long narrow beam leads is avoided, allowing fairly high values of Q to be achieved, even with normal metal plates.
While this conventional MEMS variable capacitor structure is capable of improved Q""s and avoids the intermodulation problems of varactors and xe2x80x9ctunable materialsxe2x80x9d, it has some potential problems of its own. Because only the relatively weak electrostatic attraction between plates is used to drive the plate motion to vary the capacitance, the plate support xe2x80x9cspiderxe2x80x9d structure must be extremely compliant to allow adequate motion with supportable values of bias voltage. A highly compliant suspension of even a small plate mass may render the device subject to microphonics problems (showing up as fluctuations in capacitance induced by mechanical vibrations or environmental noise). Having the electric field which drives the plates directly in the (vacuum) signal dielectric gap may cause another problem. In order to achieve a high tuning range (in this case, the ratio of the capacitance with maximum dc bias applied to that with no dc bias), the ratio of the minimum plate separation to the zero-bias plate separation must be large (e.g., 10xc3x97 would be desirable). Unfortunately, the minimum gap between the plates (maximum capacitance, and correspondingly, maximum danger of breakdown or xe2x80x9cflash-overxe2x80x9d failure between the plates) is achieved under exactly the wrong bias conditions: when the dc bias voltage is at a maximum.
Therefore, there is a need in the art for a new tunable filters incorporating MEMS-like HTS variable capacitor structure which offers extremely high Q wide tuning range, freedom from microphonics and breakdown problems, and fully isolated port tuning.
In one innovative aspect, the present invention comprises a tunable filter circuit including a variable capacitor. The variable capacitor comprises a fixed substrate on which is disposed a fixed plate(s). Preferably, the fixed plate is comprised of a high-temperature superconductor material. Adjacent to the fixed substrate is a movable substrate on which is a second plate comprised of either a normal metal or (for even higher Q) a high-temperature superconductor material. A filter signal voltage is coupled to the fixed and second plates that are disposed between the fixed and movable substrates and substantially parallel to one another such that the fixed and second plates define a capacitor gap(s) across which the filter signal voltage is impressed. A mechanical driver, operatively coupled to a tuning signal, is mounted on the fixed substrate and mounted to the movable substrate whereby mechanical displacements (for example, length displacements or other changes in shape) in the driver, in response to changes in the tuning signal, causes changes in the magnitude of the gap separating the movable plate and the fixed plate. Because the fixed and second plates define a capacitor, such gap alterations affect the capacitance(s), and in turn affect the functioning of the tunable filter circuit. Thus, a change in the tuning signal alters the frequency response of the tunable filter.
In a preferred embodiment, the fixed plate is split so that a first fixed plate and a second fixed plate both comprising a high-temperature superconductor material are on the fixed substrate. A floating plate, also comprising a high-temperature superconductor material is on the movable substrate. The fixed plates and the floating plates are disposed between the fixed and movable substrates so as to define a capacitor gap. Rather than coupled across the capacitor gap, the filter signal is coupled to the first and second fixed plates. Alternatively, the first and second fixed plates and the floating plate may comprise a normal metal (such as Ag) although such a substitution lowers the achievable Q.
The mechanical driver provides a robust, fast and well-controlled variation in the gaps(s) between the movable plate and the fixed plate(s) of the variable capacitor. The mechanical driver may provide relatively high forces over plate gap changes of at least a few microns (and preferably 10 microns). The mechanical driver may incorporate conventional piezoelectric or electrostrictive ceramic actuator materials such as PZT (lead zirconate titanate) which provide length displacements on the order of 10xe2x88x923 (the PZT actuator changes its total length approximately 0.1% in response the tuning signal). Alternatively, the mechanical driver may incorporate newer classes of materials such as higher-performance piezoelectric ceramics or electroactive polymer materials which provide a greater achievable mechanical displacement with respect to the dimensions of the mechanical driver. In response to a tuning signal, such materials have mechanical displacements that are more complex than the simple length displacements induced in a PZT actuator. However, the geometry of the mechanical driver may be altered to allow these alternate materials to be utilized.
In a preferred embodiment wherein the mechanical driver incorporates a length-displacement actuator material such as PZT, the mechanical driver comprises a folded piezoelectric or electrostrictive structure coupled to a tuning signal wherein a first piezoelectric driver is attached to the fixed substrate and to a coupling member. A second piezoelectric driver is also attached to the coupling member and to the movable substrate wherein a length decrease in the first piezoelectric driver and a length increase in the second piezoelectric driver causes the capacitor gap to decrease. The tunable filter may further comprise an HTS inductor coupled between the first and second fixed plates. Preferably, the HTS inductor has an electrical length chosen to support resonance either at the HTS inductor""s fundamental resonant frequency or at harmonics of the fundamental resonant frequency.
The present invention avoids intermodulation problems by the use of a mechanical driver to affect the capacitance because the masses involved in the mechanical tuning processes do not allow substantial motion of the capacitor plates to occur at the signal frequencies of interest. In addition, the present invention permits the tuning signal and the filter signal to be de-coupled, avoiding the prior art problems associated with two-terminal designs. The present invention offers an extremely high Q, driven in one embodiment by piezoelectric actuators for tuning high speed and low microphonics.